春天里吃大米
原文 Ultrasonic distance meter Document Type and Number:United States Patent 5442592 Abstract:An ultrasonic distance meter cancels out the effects of temperature and humidity variations by including a measuring unit and a reference unit. In each of the units, a repetitive series of pulses is generated, each having a repetition rate directly related to the respective distance between an electroacoustic transmitter and an electroacoustic receiver. The pulse trains are provided to respective counters, and the ratio of the counter outputs is utilized to determine the distance being measured. Publication Date:08/15/1995 Primary Examiner:Lobo, Ian J. 一、BACKGROUND OF THE INVENTION This invention relates to apparatus for the measurement of distance and, more particularly, to such apparatus which transmits ultrasonic waves between two points. Precision machine tools must be calibrated. In the past, this has been accomplished utilizing mechanical devices such as calipers, micrometers, and the like. However, the use of such devices does not readily lend itself to automation techniques. It is known that the distance between two points can be determined by measuring the propagation time of a wave travelling between those two points. One such type of wave is an ultrasonic, or acoustic, wave. When an ultrasonic wave travels between two points, the distance between the two points can be measured by multiplying the transit time of the wave by the wave velocity in the medium separating the two points. It is therefore an object of the present invention to provide apparatus utilizing ultrasonic waves to accurately measure the distance between two points. When the medium between the two points whose spacing is being measured is air, the sound velocity is dependent upon the temperature and humidity of the air. It is therefore a further object of the,present invention to provide apparatus of the type described which is independent of temperature and humidity variations. 二、SUMMARY OF THE INVENTION The foregoing and additional objects are attained in accordance with the principles of this invention by providing distance measuring apparatus which includes a reference unit and a measuring unit. The reference and measuring units are the same and each includes an electroacoustic transmitter and an electroacoustic receiver. The spacing between the transmitter and the receiver of the reference unit is a fixed reference distance, whereas the spacing between the transmitter and receiver of the measuring unit is the distance to be measured. In each of the units, the transmitter and receiver are coupled by a feedback loop which causes the transmitter to generate an acoustic pulse which is received by the receiver and converted into an electrical pulse which is then fed back to the transmitter, so that a repetitive series of pulses results. The repetition rate of the pulses is inversely related to the distance between the transmitter and the receiver. In each of the units, the pulses are provided to a counter. Since the reference distance is known, the ratio of the counter outputs is utilized to determine the desired distance to be measured. Since both counts are identically influenced by temperature and humidity variations, by taking the ratio of the counts, the resultant measurement becomes insensitive to such variations. 三、BRIEF DESCRIPTION OF THE DRAWINGS The foregoing will be more readily apparent upon reading the following description in conjunction with the drawing in which the single FIGURE schematically depicts apparatus constructed in accordance with the principles of this invention. 四、DETAILED DESCRIPTION Referring now to the drawing, there is shown a measuring unit 10 and a reference unit 12, both coupled to a utilization means 14. The measuring unit 10 includes an electroacoustic transmitter 16 and an electroacoustic receiver 18. The transmitter 16 includes piezoelectric material 20 sandwiched between a pair of electrodes 22 and 24. Likewise, the receiver 18 includes piezoelectric material 26 sandwiched between a pair of electrodes 28 and 30. As is known, by applying an electric field across the electrodes 22 and 24, stress is induced in the piezoelectric material 20. If the field varies, such as by the application of an electrical pulse, an acoustic wave 32 is generated. As is further known, when an acoustic wave impinges upon the receiver 18, this induces stress in the piezoelectric material 26 which causes an electrical signal to be generated across the electrodes 28 and 30. Although piezoelectric transducers have been illustrated, other electroacoustic devices may be utilized, such as, for example, electrostatic, electret or electromagnetic types. As shown, the electrodes 28 and 30 of the receiver 18 are coupled to the input of an amplifier 34, whose output is coupled to the input of a detector 36. The detector 36 is arranged to provide a signal to the pulse former 38 when the output from the amplifier 34 exceeds a predetermined level. The pulse former 38 then generates a trigger pulse which is provided to the pulse generator 40. In order to enhance the sensitivity of the system, the transducers 16 and 18 are resonantly excited. There is accordingly provided a continuous wave oscillator 42 which provides a continuous oscillating signal at a fixed frequency, preferably the resonant frequency of the transducers 16 and 18. This oscillating signal is provided to the modulator 44. To effectively excite the transmitter 16, it is preferable to provide several cycles of the resonant frequency signal, rather than a single pulse or single cycle. Accordingly, the pulse generator 40 is arranged, in response to the application thereto of a trigger pulse, to provide a control pulse to the modulator 44 having a time duration equal the time duration of a predetermined number of cycles of the oscillating signal from the oscillator 42. This control pulse causes the modulator 44 to pass a "burst" of cycles to excite the transmitter 16. When electric power is applied to the described circuitry, there is sufficient noise at the input to the amplifier 34 that its output triggers the pulse generator 40 to cause a burst of oscillating cycles to be provided across the electrodes 22 and 24 of the transmitter 16. The transmitter 16 accordingly generates an acoustic wave 32 which impinges upon the receiver 18. The receiver 18 then generates an electrical pulse which is applied to the input of the amplifier 34, which again causes triggering of the pulse generator 40. This cycle repeats itself so that a repetitive series of trigger pulses results at the output of the pulse former 38. This pulse train is applied to the counter 46, as well as to the pulse generator 40. The transmitter 16 and the receiver 18 are spaced apart by the distance "D" which it is desired to measure. The propagation time "t" for an acoustic wave 32 travelling between the transmitter 16 and the receiver 18 is given by: t=D/V s where V s is the velocity of sound in the air between the transmitter 16 and the receiver 18. The counter 46 measures the repetition rate of the trigger pulses, which is equal to 1/t. Therefore, the repetition rate is equal to V s /D. The velocity of sound in air is a function of the temperature and humidity of the air, as follows: ##EQU1## where T is the temperature, p is the partial pressure of the water vapor, H is the barometric pressure, Γ w and Γ a are the ratio of constant pressure specific heat to constant volume specific heat for water vapor and dry air, respectively. Thus, although the repetition rate of the trigger pulses is measured very accurately by the counter 46, the sound velocity is influenced by temperature and humidity so that the measured distance D cannot be determined accurately. In accordance with the principles of this invention, a reference unit 12 is provided. The reference unit 12 is of the same construction as the measuring unit 10 and therefore includes an electroacoustic transmitter 50 which includes piezoelectric material 52 sandwiched between a pair of electrodes 54 and 56, and an electroacoustic receiver 58 which includes piezoelectric material 60 sandwiched between a pair of electrodes 62 and 64. Again, transducers other than the piezoelectric type can be utilized. The transmitter 50 and the receiver 58 are spaced apart a known and fixed reference distance "D R ". The electrodes 62 and 64 are coupled to the input of the amplifier 66, whose output is coupled to the input of the detector 68. The output of the detector 68 is coupled to the pulse former 70 which generates trigger pulses. The trigger pulses are applied to the pulse generator 72 which controls the modulator 74 to pass bursts from the continuous wave oscillator 76 to the transmitter 50. The trigger pulses from the pulse former 70 are also applied to the counter 78. Preferably, all of the transducers 16, 18, 50 and 58 have the same resonant frequency. Therefore, the oscillators 42 and 76 both operate at that frequency and the pulse generators 40 and 72 provide equal width output pulses. In usage, the measuring unit 10 and the reference unit 12 are in close proximity so that the sound velocity in both of the units is the same. Although the repetition rates of the pulses in the measuring unit 10 and the reference unit 12 are each temperature and humidity dependent, it can be shown that the distance D to be measured is related to the reference distance D R as follows: i D=D R (1/t R )/(1/t) where t R is the propagation time over the distance D R in the reference unit 12. This relationship is independent of both temperature and humidity. Thus, the outputs of the counters 46 and 78 are provided as inputs to the microprocessor 90 in the utilization means 14. The microprocessor 90 is appropriately programmed to provide an output which is proportional to the ratio of the outputs of the counters 46 and 78, which in turn are proportional to the repetition rates of the respective trigger pulse trains of the measuring unit 10 and the reference unit 12. As described, this ratio is independent of temperature and humidity and, since the reference distance D R is known, provides an accurate representation of the distance D. The utilization means 14 further includes a display 92 which is coupled to and controlled by the microprocessor 90 so that an operator can readily determine the distance D. Experiments have shown that when the distance between the transmitting and receiving transducers is too small, reflections of the acoustic wave at the transducer surfaces has a not insignificant effect which degrades the measurement accuracy. Accordingly, it is preferred that each transducer pair be separated by at least a certain minimum distance, preferably about four inches. Accordingly, there has been disclosed improved apparatus for the measurement of distance utilizing ultrasonic waves. While an illustrative embodiment of the present invention has been disclosed herein, it is understood that various modifications and adaptations to the disclosed embodiment will be apparent to those of ordinary skill in the art and it is intended that this invention be limited only by the scope of the appended claims. 译文 超声波测距仪 文件类型和数目：美国专利5442592 摘要：提出了一种超声波测距仪来抵消的影响温度和湿度的变化，包括测量单元和参考资料。在每一个单位，重复的一系列脉冲的产生，每有一个重复率，直接关系到各自之间的距离，发射机和接收机。脉冲提供给各自的主机，和比例的反产出是利用确定的距离被衡量的。 出版日期： 1995年8月15日 主审查员：罗保.伊恩j. 一、背景发明 本发明涉及到仪器的测量距离，更特别是，这种仪器传送超声波两点之间。 精密机床必须校准。在过去，这已经完成利用机械设备，如卡钳，微米等。不过，使用这种装置并不容易本身自动化技术。据了解，该两点之间距离才能确定通过测量传播时间的浪潮往返那些两点。这样一个类型的波是一种超声波，或声，海浪。当超声波旅行两点之间，距离两个点之间可以衡量乘以过境的时间波由波速，在中期分开两点。因此，这是一个对象本发明提供仪器利用超声波准确测量两点之间距离。 当中等两个点之间的间距是被衡量的是空气，声速是取决于温度和空气相对湿度。因此，它是进一步对象的，现在的发明，提供仪器的类型所描述的是独立于温度和湿度的变化。 二、综述发明 前述的和额外的对象是达到了根据这些原则的这项发明提供距离测量仪器，其中包括一个参考的单位和测量单位。参考和测量单位是相同的，每个包括一电发射机和接收机一电。间隔发射器和接收器的参考股是一个固定的参考距离，而间距之间的发射机和接收机的测量单位是距离来衡量。在每一个单位，发射机和接收机是再加上由一个反馈环路导致发射机产生的声脉冲是由接收机和转换成一个电脉冲这是然后反馈到发射机，使重复一系列脉冲的结果。重复率脉冲是成反比关系之间的距离发射器和接收器。在每一个单位，脉冲提供一个反。由于参考的距离是众所周知，比例反产出是利用，以确定所期望的距离来衡量。由于这两方面都是相同的影响，温度和湿度的变化，采取的比例罪状，由此产生的测量变得麻木等变化。 三、简要说明图纸 前述将更加明显后，读下列的说明，在与该绘图并在其中单一数字schematically描绘仪器兴建根据这些原则的这项发明。 四、详细说明 谈到现在的绘图，有结果表明，测量单位和10个参考单位12个，均加上一个利用的手段14 。测量单位包括1 10电发射机16日和1电接收机18 。变送器16包括压电材料20夹心阶层之间的对电极的22日和24日。同样，接收机18个，包括压电材料26夹心阶层之间的对电极的28日和30日。作为众所周知，采用电场整个电极22日和24日，强调的是，诱导，在压电材料20 。如果该字段各有不同，如所申请的一个电脉冲，声波是32所产生的。为进一步众所周知，当声波影响到接收器18 ，这诱导应力，在压电材料26 ，导致一种电信号，以产生全国电极28日和30日。虽然压电传感器已说明，其他电声装置，可利用，例如，静电，驻极体或电磁类型。 如表所示，电极28日和30日的接收18岁以下的耦合的投入一34放大器，其输出耦合输入一个探测器36 。探测器36是安排提供一个信号，脉冲前38时，输出放大器34已经超过预定的水平。脉冲前38 ，然后产生一个触发脉冲，这是提供给脉冲发生器40 。在为了提高灵敏度，该系统，传感器16和18岁以下的共振兴奋。有相应的提供了一个连续波振荡器42提供了一个连续振荡信号在一个固定的频率，最好是共振频率的传感器16和18 。这个振荡信号是提供给调制器44 。要有效地激发发射机16 ，可取的做法是提供几个周期的共振频率信号，而不是一个单脉冲或单周期。因此，脉冲发生器40是安排，在回应的应用存在的一个触发脉冲，提供一个控制脉冲调制器44有一个时间的平等的时间，时间预定人数的周期振荡信号从振荡器42 。这个控制脉冲调制器的原因， 44个通过了“水管爆裂”的周期，以激发发射机16 。 当电力是适用于所描述的电路，有足够的噪音在输入到放大器34 ，其输出触发脉冲发生器40至造成了一片叫好声，振荡周期，以提供整个电极22日和24日的发射器16 。变送器16因此产生声波32条，其中影响到接收器18 。接收器18 ，然后产生一个电脉冲，这是适用于输入放大器的34 ，这再次触发原因的脉冲发生器40 。这个周期重演，使重复一系列的触发脉冲结果的输出脉冲前38 。这脉冲列车是应用到46个柜位，以及向脉冲发生器40 。 变送器16日和接收18岁以下的间隔，除了由距离的“ D ” ，它是理想的衡量。传播时间的“ T ”为一声波32往来变送器16日和接收18所给予的： = D的吨/视频s 凡v s是声速在空气中之间的发射机16日和接收18 。柜台46措施重复率触发脉冲，这是平等的1 /汤匙因此，重复率是平等的一至中五的S /四该声速空气中是一个功能的温度和湿度的空气，内容如下： ＃ ＃ ＃ ＃ equ1其中T是温度， P是局部的压力，水汽， H是该气压， γ瓦特和γ一顷的比例不断的压力，具体的热不断货量具体的热水汽和干燥的空气，分别。因此，虽然重复率触发脉冲测量非常准确地反46 ，声速的影响，温度和湿度，使测量的距离d无法确定准确。 根据这些原则的这项发明，参考单位提供的是12 。参考单位12是相同的建设为测量单位的10个，因此，包括一电发射机50个，其中包括压电材料52夹心之间的一对电极的54和56 ，和一电接收机58 ，其中包括压电材料60夹心阶层之间的一对电极60,61,62和64 。再次，传感器以外的其他类型压电可以利用。变送器50和接收五十八顷间隔，除了已知的和固定的参考距离“博士” 。电极60,61,62和64耦合到输入的放大器66 ，其输出是耦合的投入探测器68 。输出探测器68是耦合的脉搏，前70产生触发脉冲。触发脉冲应用到脉冲发生器的72个控制调制器74通过扫射从连续波振荡器76至变送器50 。触发脉冲从脉冲前70也适用于反78 。 最好是，所有的传感器16 ， 18 ， 50和58具有相同的共振频率。因此，振荡器42和76都在运作，频率和脉冲发电机40和第72条提供平等的输出脉冲宽度。 在用法上，测量装置10和参考资料股一十二顷在接近，使该声速在这两个单位是相同的。虽然留级率的脉冲在测量单位， 10和参考资料股十二顷每个温度和湿度的依赖性，能证明的距离D来衡量。 其中T R是传播时间超过距离博士在参考股12 。这种关系是独立于双方的温度和湿度。 因此，产出的柜台46和78所提供的投入微处理器的90个利用的手段14 。微处理器90是适当的程序提供了一个输出是成正比的比例，产出的柜台46和78 ，这反过来又是成正比的重复率分别触发脉冲列车的测量单位， 10和参考资料股12 。作为描述，这个比例是独立的温度和湿度，由于参考的距离，博士，是众所周知的，提供了一个准确的代表性距离四，利用手段， 14日还包括一个显示92这是耦合和控制的微处理器，使90一个经营者可以随时确定的距离四 实验表明，当之间的距离发射和接收传感器是太小了，思考的声波在传感器的表面有一个不小的作用，降低了测量精度。因此，最好是每换一双分开，至少由某一个最小距离，最好是约四英寸。 因此，已披露的改善仪器的测量距离，利用超声波。而一个说明性的体现，本发明已披露者外，据了解，各种修改和适应所披露的体现，将是显而易见的那些普通的技巧与艺术，这是打算把这个发明只限于由范围所附的索赔。
chenmingzhu
从天线接收的微弱信号由处于射频接收机前端的放大器进行放大，因此要求该放大器具有一定的增益和较小的噪声系数。 本文借助Agilent公司的射频电路设计软件ADS(Advanced Design System)进行辅助设计一款高增益低噪声放大器(LNA)，并对其进行了仿真验证。1 射频放大器的组成 单级射频放大器的组成如图1所示，包括射频晶体管放大电路和输入、输出匹配网络三部分。2 射频放大器的设计2．1 晶体管的选择 选择好晶体管器件对低噪声放大器的设计至关重要。 根据工作频率、增益和噪声系数等指标要求，同时考虑到设计、仿真时便于得到相应的元器件模型，最终选用Avago公司的高电子迁移率晶体管(E-PHEMT)ATF-58143来进行设计(可以在Avago公司的网站上下载到ATF-58143的元件模型)。2．2 偏置电路的设计 设计LNA首先需要确定静态工作点，利用ADS中的“DC_FET_T”的模板可以很方便地仿真出其输出特性曲线。再参考ATF-58143的datash eet，可以确定当Vds=3 V，Ids=35 mA时，各项设计指标满足要求。 确定静态工作点后，就要确定偏置电路的形式和参数。不需人工计算，借助ADS中的设计向导工具(DesignGuide→Amplifier→Tools→ Transistor Bias Utility)可以轻易完成。因为ADS所提供的元件数值是非标称的，所以需要设计者用与ADS提供的数值接近的标称元件进行替代。偏置电路及各点静态参数如图2所示。2．3 稳定性分析及改善 晶体管绝对稳定的条件是K>1，|△|<1。其中： 如果这两个条件不能同时得到满足，电路将存在潜在的不稳定和振荡的可能。对上述偏置条件下的晶体管进行稳定性仿真分析发现，在要求的工作频段内其稳定系数K<1，不满足绝对稳定的条件。 通过引入负反馈的方式可以改善电路的稳定性，同时也能够拓展工作带宽。在输出端和输入端之间串联RC电路引入负反馈，其中的R需要满足条件： 同时在两个源极加上小的电感引入负反馈进一步改善稳定性，该电感的值需反复调节后方能确定。 对引入负反馈后的电路再次仿真，其工作频带内稳定系数K>1，满足绝对稳定条件。2．4 最小噪声系数的输入匹配电路设计，最大增益的输出匹配电路设计 如果输入匹配电路和输出匹配电路使射频器件的输入阻抗Zin和输出阻抗Zout都转换到标准系统阻抗Zo，即Zin=Zo，Zout=Zo(或，如图1所示)就可使器件的传输增益最高。但输入、输出匹配时，噪声并非最佳。当ΓS=Γopt时，可以得最小的噪声系数。 利用ADS可以很方便地绘制出等功率增益圆和等噪声系数圆，如图3所示。从图中可以看出，如果从m2点匹配到标准系统阻抗，将可以使电路获得最大的增益；如果从m3点匹配到标准系统阻抗，将可获得最小的噪声系数。显然最大增益和最小噪声系数不可同时得到。对于低噪声放大器，首要的是考虑最小噪声系数，因此对m3点进行匹配。借用ADS的自带工具“Smith Chart Utility Tool”进行，只要在其中设置好频率、源阻抗和目标阻抗值，就可以设计出所需要的输入匹配电路。 在输入端匹配完成以后，在原理图中加入阻抗测量控件测出输出阻抗，再次使用“Smith Chart Utility Tool”将输出阻抗匹配到标准系统阻抗，就可得到最大增益的输出匹配电路。 当输出端的匹配完成后，因为改变了从输入端向里看的等效阻抗Zin，输入端的回波损耗会变差。为此，可以采用优化控件对输入端和输出端的匹配电路进行同时的优化改进，也可以使用Tunig工具进行调节。2．5 最终电路及仿真结果分析 匹配及优化后的电路如图4所示，电路中各元件的作用分别是：C6、L6是输入匹配电路；C7、L7是输出匹配电路；L1、L5、C3、R5是反馈元件；L3、L4是扼流电感；C4、C5是隔直耦合电容；C1、C2是旁路电容。 需要说明的是，反馈电感L1、L5和匹配电路中的元件C6、L6、C7、L7等因为数值较小，在工程中常用微带线来代替。 仿真结果如图5所示。其工作带宽达500 MHz，中心频率处增益接近20 dB，输入输出反射损耗小于-10 dB，噪声系数小于0．5 dB，稳定系数大于1。如果断开反馈电路后再次仿真，会发现增益有所加大，但稳定系数将小于1，放大电路将不能正常工作。3 结论 通过射频低噪声放大器的设计与仿真，可以看到使用ADS辅助设计电路，理论计算简单，设计过程快速，参数修改容易，验证方便，缩短了设计周期，提高了设计精度，在工程中具有实用价值。
By a weak signal from the antenna in the rf receiver front-end amplifier amplification, therefore asked the amplifier gain and low noise factor.
In this paper, with the aid of Agilent ADS of rf circuit Design software (Advanced Design System) for aided Design a high gain and low noise amplifier (LNA), and the simulation verification.
1 the composition of the rf amplifier
Single stage composed of rf amplifier is shown in figure 1, including rf transistor amplifier circuit and the input and output matching network of three parts.
2 the design of the rf amplifier
2.1 the choice of the transistor
Good selection transistor components for the design of the low noise amplifier is very important.
According to the working frequency, gain and noise figure index requirements, at the same time when considering the design, the simulation is easy to get the corresponding components model, finally choose Avago company of high electron mobility transistor (PHEMT) E ATF - 58143 for design (can be downloaded on Avago company web site to the ATF components model - 58143).
2.2 the design of the bias circuit
Designing LNA first need to determine the static working point, the use of ADS "DC_FET_T" templates can be easily in the simulation of the output characteristic curve. Reference ATF - 58143 again datash eet, can be determined when the Vds = 3 V, Ids = 35 mA, the design indexes meet the requirements.
After determine the static working point, shall determine the form and the parameters of bias circuit. Do not need artificial calculation, with the aid of ADS in the design wizard tool (DesignGuide - Amplifier - > Tools - Transistor Bias, the Utility) can be done easily. Because the ADS provided by the component values are nominal, so designers need to use with the ADS provide alternative values close to the nominal elements. Bias circuit and some static parameters as shown in figure 2.
2.3 stability analysis and improvement
Transistor is K > 1, the absolute and stability of the | delta | < 1. Among them:
If the two conditions cannot be satisfied at the same time, there will be potential instability and oscillatory circuit. Transistor of the bias conditions stability simulation analysis found that the stability coefficient within the required working frequency band K < 1, can not meet the needs of absolute stability conditions.
By introducing feedback on ways to improve the stability of the circuit, but also can extend working bandwidth. Between the output and the input series RC circuit is introduced into feedback, of which R need to meet the conditions:
In both the source and small inductance is introduced into feedback to further improve the stability, the value of the inductance to repeatedly adjust the rear can be determined.
Introduction of negative feedback circuit simulation again, within its working frequency stability factor K > 1, meet the absolute stability condition.
2.4 minimum noise factor input matching circuit is designed, the biggest gain of the output matching circuit design
If the input matching circuit and the output matching circuit of rf devices Zin the input impedance and output impedance Zout impedance Zo are transformed to the standard system, namely the Zin = Zo, Zout = Zo (or, as shown in figure 1) to make a device transport the highest gain. But when input and output matching, noise is not the best. When Γ S = Γ opt, could get the minimum noise figure.
ADS can be easily draw power gain and noise coefficient, as shown in figure 3. Can be seen from the diagram, if from m2 point impedance matching to the standard system, will be able to make the circuit gain maximum gain; If impedance matching to the standard system, from the m3 point will be minimal noise coefficient can be obtained. Obviously the biggest gain and the minimum noise figure cannot get at the same time. For low noise amplifier, the first is to consider the minimum noise figure, and so on m3 point matching. Use ADS bring tools "Smith Chart the Utility Tool", in which as long as the set frequency, source impedance and the target impedance value, can the input matching circuit design need.
In the input matching is complete, add impedance measurement control measure in principle diagram output impedance, again using "Smith Chart the Utility Tool will impedance, output impedance matching to the standard system can get the maximum gain of the output matching circuit.
When the output matching is completed, because has changed from the input to see the equivalent impedance Zin, will get poor return loss at the input. For this purpose, the optimal control can be used for the input and the output matching circuit optimization to improve at the same time, also can use Tunig tools.
2.5 the final circuit analysis and simulation results
Matched and optimized circuit as shown in figure 4, the role of each element in the circuit are respectively: C6, L6 is input matching circuit; C7, is about the output matching circuit; L1, L5, C3, R5 is feedback element; L3, L4 is choke inductance; C4, C5 is the direct coupling capacitance; C1, C2 is the bypass capacitor.
Feedback to be sure, inductance L1, L5 and matching circuit element in C6, L6, C7, about because small amounts, such as microstrip line to replace the commonly used in engineering.
The simulation results as shown in figure 5. Its working bandwidth of 500 MHz, the center frequency close to 20 dB gain, input and output return loss is less than 10 dB of noise coefficient is less than 0.5 dB, stability factor greater than 1. If disconnect again after feedback circuit simulation, will find the gain increased, but the stability coefficient will be less than 1, the amplifying circuit will not work properly.
3 conclusion
Through radio frequency low noise amplifier design and simulation, can see use ADS auxiliary circuit design, the theoretical calculation is simple, rapid design process, parameter modification easy, convenient, shorten the design cycle,
